Simple two-electrode biosignal amplifier

Simple two-electrode biosignal amplifier D. Dobrev T. Neycheva N. Mudrov 1Centre of Biomedical Engineering 'Ivan Daskalov', Bulgarian Academy of Sciences, Sofia, Bulgaria Abstract--A simple, cost effective circuit for a two-electrode non-differential biopotential amplifier is proposed. It uses a "virtual ground" transimpedance amplifier and a parallel RC network for input common mode current equalisation, while the signal input impedance preserves its high value. With this innovative interface circuit, a simple non-inverting amplifier fully emulates high CMRR differential. The amplifier equivalent CMRR (typical range from 70-100 dB) is equal to the open loop gain of the operational amplifier used in the transimpedance interface stage. The circuit has very simple structure and utilises a small number of popular components. The amplifier is intended for u s e in various two-electrode applications, such as Holter-type monitors, defibrillators, ECG monitors, biotelemetry devices etc. Keywords--ECG amplifier, Biopotential amplifier, Non-differential amplifier, Common mode interference Med. Biol. Eng. Comput., 2005, 43, 725-730 1 Introduction THE USE of conventional, unsymmetrical amplifier circuits in biomedical engineering is very limited, because of their inadequacy in suppressing power-line interference. Because one of the patient electrodes is directly connected to the amplifier signal ground and the other is a high impedance point, the interference current flows only through the grounded electrode. The grounded electrode impedance voltage drop is amplified and leads to circuit saturation or masking of the useful biopotential signal. Many biosignal acquisition devices could benefit from the use of only two electrodes. Electrocardiogram monitoring in intensive care wards, ambulatory monitors, defibrillators etc. are among the most common examples. The most widely used technique for biosignal amplification is based on an instrumentation amplifier first-stage design, because of its ability to suppress the common mode interference (NEUMAN, 1998). In two-electrode instrumentation, the amplifier inputs must have a differential impedance that is as high as possible to avoid signal attenuation. On the other hand, the amplifier should have reasonably low common mode impedance, to create a path for the common mode interference currents without significant voltage drop, keeping both inputs in their specified operating voltage range. THAKOR and WEBSTER (1980) introduced a bootstrapped input stage. However, it has an inductive common mode input impedance, resulting in a very poor common mode input current range, especially for higher frequencies. A circuit for a two-electrode, non-differential amplifier was developed (DOBREV, 2002), the performance of which is Correspondence should be addressed to Dr. Dobromir Dobrew emaih dobri@clbme.bas.bg Paper received 22 February 2005 and in final form 30 September 2005 MBEConline number: 20054061 9 IFMBE: 2005 Medical & Biological Engineering & Computing 2005, Vol. 43 quasi-equivalent to the differential amplifier described by DOBREV and DASKALOV (2002). Because the body as signal source is floating and, in addition, the modern biopotential amplifiers are isolated, it is possible to drive actively one of the electrodes to the circuit common potential and thus to balance the flowing interference currents. Now, a very simple, low-cost circuit of a two-electrode nondifferential amplifier is suggested, where the interference current balancing is achieved by one operational amplifier (OA) and two parallel RC networks. 2 Body-amplifier interface circuit concept An equivalent circuit of the body-amplifier interface is presented in Fig.1. The interference current Ipi (200 nA typical value; can reach a few microamperes in close approach to power-line connected cables or instruments) is mainly defined by the power-line-body stray capacitance Cp: Ipi o)Cpgpl, because Cp < Cu < Cb and l/toCp >> R b a , Rbb, Zea, Zeb, ZFea, ZFeb. The skin-electrode impedances are Zea and Z~b, and Cu is the capacitance between the circuit ground and earth. A part of this interference current Cb Icb = Ipl Cb + Cg flows through the body impedances Rba, Rbb and the stray capacitance to earth Cb. Another part of the interference current c~ /g = / . +/b =/pZ cb + cg traverses the impedances Zea (Ia) , Zeb(Ib) , the output of OA A1, which is a grounded voltage source, and C~ to earth. 725 J Rba Zea Ia T,. :lw.<.c, 2~ r " u? l '----./ ] 7 +.J/ As the OA A 1 maintains the potential at input b equal to its offset voltage (virtual ground), A2 amplifies the voltage between inputs a and b. It is interesting to establish how stable the 'virtual ground' voltage is at node b and what the equivalent CMRR of the circuit is if we compare it with the differential one with the same input impedances ZIN, = ZFea, ZINb = Zmb, the same gain and the same output level of interference. For a differential amplifier, the common mode input error is WcmAcm Wcm e err - - _ _ - - _ _ Ad Fig. 1 CMRR CMRR CMRR Equivalent circuit of patient-amplifier interface The input currents I, and Ib have common mode Icm and differential mode Ia components (DOBREV and DASKALOV, 2002) /.+/b 2 Icm----, where A .... and Aa are the amplifier common and differential mode gain coefficients, and Zorn is the amplifier common mode impedance component per input. For the proposed circuit, the voltage variation at node b is the interface circuit error Ij=I.--Ib eerr = Vb =[cmZb =[cm - ZFBb ~ ' Alo~ + 1 or Ia = I c m § 2 4 7 Ib =Icm 2 --Icm--Iecg Here, Iec~ = Ia/2 is the current generated by the body as biopotential signal source Ve~- Note that Ic,, has the same values and directions for both inputs, whereas Ie~ has equal values, but is in opposite directions. The input current Ib is converted to voltage at the output of the OA A 1 t h r o u g h the feedback impedance Zmb. A1 drives and equalises the input b potential to the common point. In addition, A 1 o u t p u t is connected to the amplifier input a by impedance Zma. The potential at node a is a superposition from Zma and Zmb voltage drops achieved by flowing common mode interference (Ic,,) and differential mode signal (Ie~) input currents. In the following consideration, the body resistances Rba and Rbb continue to be neglected, as they have a very small value compared with the other impedances. For common mode interference, the remaining impedances Zea, Zeb, Zma, Zmb take the shape of the familiar Wheatstone bridge. If the ratio Zea to ZFB a is precisely the same as the ratio Zeb to Zmb, the bridge is 'nullified' or balanced Z~ [cmZFBb - Alo~ Here Zb is the 'virtual ground' impedance at node b, and Alo L is the A 1 open loop gain. If we compare eerr in the two equations above, it can be seen that the equivalent CMRR of the described circuit is equal to the open loop gain of the OA A 1 CMRReqv = AloL This conclusion is a general rule of thumb when selecting the OA A 1. The selection should be done as follows: First, because of high-ohmic resistors used in the circuit, all OAs should have a MOSFET or JFET input stage to minimise the offset voltage error created by their input bias currents. Next, the OA A 1 should be unity-gain stable, because it works as a current to voltage converter. Finally, using the parameter specified in the OA datasheet gain bandwidth product (GBP) (or unity gain bandwidth) and considering first-order open loop gain roll-off, the CMRReq~for an arbitrary frequency of interest can be easily calculated CMRReqv(f) = 20 log G f PAl Z~b ZFBa--ZFB b In this ideal condition, no interference signal will appear as voltage drop between points a and b. Considering that, the input current common mode component Ic,, has the same value for I, and, Ib, and, if Zma is identical to Zmb, then the corresponding impedance voltage drops axe equal, and the circuit becomes quasi-symmetric with respect to the interference. Assuming that Zea equalled Zeb, the above bridge equation would be met, and the interference would be fully cancelled. However, in real life, Zea and Zeb are not equal, and the flowing common mode interference current Ic,, multiplied by their difference AZe will cause an unwanted residual interference signal at node a Va = AZelcm = (Zea -- Zeb) Ia § Ib 2 amplified by A2. 726 CMRR [cmZcm [cmZ1Na [cmZ1Nb -- _ _ -- _ _ - - _ _ For example, if an OA with GBP = 3 MHz, e.g. TL082, MCP602, is selected, then CMRReqv(50 H z ) = 95 dB. If low power and a slow OA, e.g. MCP607 with GBP = 150 kHz are used, then CMRReqv(50Hz) = 69 dB. Note that CMRReq~, calculated in such a way, cannot exceed the OA open loop gain DC value specified in the datasheet. As the signal source Vecs generates only differential currents (in opposite direction for both inputs), the amplifier input impedance (between inputs a and b) is ZIN = Zma + Zmb. The common mode input impedance between both inputs and ground has a very low value ZCM -- ZFsblIZFSa AloL + 1 Zb 2 The amplifier common mode current capability (both inputs) IcM,,ox depends on Zma, Zmb, the selected supply voltage Vcc and the A 1 o u t p u t stage voltage drop Vos at maximum Medical & Biological Engineering & Computing 2005, Vol. 43 signal swing ICMmax - - Vcc - Vos 2(Vcc - Vos) -ZFB~IIZFBb ZFB 2Vcc ~' - - ZFB The Ic~m,x value should be at least 1 ~ A for batterysupplied telemetric or Holter-type devices installed close to the body surface, and at least 5 - 6 IxA for monitors with conventional leads 9 3 Practical circuits and design tips A very low cost circuit of an ECG amplifier built according to the proposed principle, is shown in Fig.2. It is powered by a _+5 V split supply voltage source. The low cost and very popular general purpose JEET OAs TL082 (TL084) are selected 9The amplifier performances axe listed below: 9 frequency band: 0 . 0 5 - 1 0 0 Hz 9 gain: 525 (54.4 dB) 9 input impedance Z I N = 2(Rprt + RFB/1 + SCFBRFB) 9 input DC differential voltage range: 0.33 V 9 input common mode current range: ICMmax -- 2(Vcc - Vos) ZFB ~. . /(~oCFsRFs) 2 + 1 = 2(Vcc - ZVbe)f For TL082, the output stage voltage drop at maximum signal swing is about two base-emitter junctions: Vos = 2Vb~ therefore Icmm~(50 Hz) ~ 5.4 ~A 9 transfer function: A(s) = RFB 1 RF8 + Rprt 1 § SCFB(RFBIIRprt) R1 + R2 1 + sCI(RIIIR2) sC2R3 R2 1 + sC1R1 1 + sC2R3 R4 § R5 1 + sC3(R411Rs) R5 1 § sC3R4 Protective input resistors Rp~ axe inserted and used for RF noise filtering also 9 The value of the time constant CFBRp~t can be derived from the following consideration 9 CfB(RfBllRprt) = C3(R4IIRs) ~ CFBRprt = C3R5, the With second high frequency zero in the amplifier transfer function is cancelled 9 The following practical recommendations should be taken into account 9 The protective resistor values Rp~t should be selected in range of 1 0 - 4 7 k. Higher values are not convenient, as the divider with RFB would reduce the biosignal amplitude and the signal-to-noise ratio 9Lower values would weaken the protection against radio-frequency interference 9 CFB is to be selected in view of the following limitations 9 A very small CFe would reduce the amplifier stability, RF noise filtering and input common mode current range, CFe > 2.2 nF would reduce the input impedance for the signal frequency band. Therefore a recommendable range for CFe values is 330 p F - 2 . 2 nF. RFe can be selected from 2 . 2 - 1 0 M ~ , depending on the desired input impedance, supply voltage used and expected level of interference 9RFB < 2.2 M ~ would reduce the amplifier input impedance 9A higher RFe value would degrade the input common mode current range, possibly leading to saturation of A1, especially when CFe is small and the supply voltage is low. All resistors used in the circuit have 1% tolerance. The capacitors CFe axe NP0 type, with 5% tolerance 9All remaining capacitors axe X7R type, with 10% tolerance 9 The component cost of the described amplifier circuit is less than $0.50 (see the Avnet price list http://www.em.avnet.com). It is important to avoid an additional low-resistance DC link between a point on the body and the circuit ground, for example by a third electrode 9Such a condition would cause amplification of the input offset voltage of A1 and is able to block the next stage 9Therefore this circuit is intended only for use with two electrodes 9A three-electrode connection can be used, if deemed necessary 9The third electrode should be connected through a ceramic (non-electrolytic) capacitor of, for example, 10 ~F, Y5V type, and Rprt should be increased to 4 7 - 6 8 k ~ . A practical ECG circuit for 5 V single supply operation is shown in. It is rather complicated, with increased low-pass filtering and added protection against defibrillator shocks 9 Because two dual (or one quad) OA ICs are enough for the signal amplification, the fourth, spare OA can be used to provide a low impedance ground reference 9 The low-pass RC network, inserted after Uza with time constant CIR2, cancels the first high-frequency zero in the amplifier transfer function 9The time constant CFe(RDF + Rpr~) is chosen three times bigger than C3R5 and ensures the second high frequency zero effect elimination 9Also, additional RF noise filtering is implemented by the time constant CRFRp~ (for common mode currents). As the OA input differential pair can act as an RF detector separating a lower-frequency Vcc U1s Rprt l 10 k CFB- BAV99 VEE VCC Rprt CFB 2.2n U~A 2.2n RFB R1 100k R3 47:: 3.3M ,oR: 3.3M 10k VEE Fig. 2 Practical, very low-cost EGG amplifier Medical & Biological Engineering & Computing 2005, Vol. 43 727 5V RDF Rprt DF MCP607 I BAV99 RFB 2.2M Crf BZV55C10 _~ ~- 33C0p _ CUB MCP602 R2 10n T lOn 1C0'Irl Rprt zvo c,o: 22;~ DDF Crf MCP607 RI RFB RDF C2 C3 33RM I~U BAV9 2 U1B ~ 1M ~ 10~ = Fig. 3 Practical single supplied ECG amplifier with additional filtering and defibrillator protection envelope from a carrier, the RF interference can simply increase the OA input offset voltage or even transfer an amplified envelope replica to the circuit output. The added capacitors CRF additionally protect U2e and UIA from incoming R F noise. The inserted resistors RDF and Zeners diodes DDF protect against defibrillator shocks. Their values should be defined by the following consideration. The maximum energy delivered into the body by the present-day defibrillators is limited to 360 J, owing to possible myocardial cell damage. The required energy is achieved usually by a 47 ixF capacitor charged to 4000 V. At defibrillation time, the body resistance Rb.,.x can have a maximum value of 200 ~ . Rb.,.x and both protection resistors RDF axe in parallel and implement a current divider, therefore the energy absorbed by one RDF resistor is half of all delivered energy multiplied by the current divider gain 360J Rbmax JRDF = ~ - - " Rb,~x + 2RDF -- 0.46J A good choice for RDF is the OhmCraft high-voltage resistor series (HVC series, Data Sheet), in the SMD 2512 package, which can absorb up to 2 J energy without failure, has an operating range of 2500 V and can withstand up to 5000 V surcharge. The Zener diodes DDF should be selected in the 8 . 2 - 1 2 V range. These values minimise the effects of higher leakage channelN 0-~ channel1 current and obtuse knee characteristic related to the lowvoltage Zener diodies. The 0.5 W Minimelf package for DDF is enough. The circuit described can be easily extended to a multichannel configuration by simple reproduction, except for the transimpedance stage, which is common for all channels, as shown in Fig. 4. The parallel RC networks used must be equal. They can be implemented with a slightly higher impedance value, e.g. 10 M O 11330 pF. 4 Results and conclusions Two ECG signal records acquired using the circuit illustrated in Fig. 2 and a differential two-electrode ECG amplifier with AD620 input stage are shown in Figs 5 and 6. The AD620 input stage replaces the functions of the stages around UIA and UI~ in Fig. 2. It is set with a gain of 11 (like the UI~ stage) and uses exactly the same passive components: resistors Rpr~ connected to AD620 inputs, and RFB, CFB connected between each AD620 input and the circuit ground. The breadboard is supplied by two 9 V batteries and is connected with nonshielded wires, of approximately 80 cm length, to a pair of electrodes located on the chest (modified lead I). Only one input stage at a time can be selected by jumpers. Looking at Figs 5 and 6, it is clear that the power-line interference levels in both circuits are almost the same. The first ~ iI~l i [!lI filI i,ili Q Fig. 4 728 Ground-free multichannel amplifier principle Fig. 5 Electrocardiogram acquired with AD620 differential input stage amplifier Medical & Biological Engineering & Computing 2005, Vol. 43 imbalance). The measured interference level in the ECG signal was approximately the same and the original, CFB capacitors were restored. The CFB capacitors were measured, and a 40 pF (2%) imbalance was found. Next, the input resistor pairs Rprt and RFB were measured, and an imbalance of 20 (0.2%) for Rprt and 1 0 k f l (0.3%) for RFB was found. Both inputs of the circuit were connected by a 22 IxF tantalum capacitor to a 50 Hz sinusoidal voltage generator. The generator amplitude was set to 2 mV to generate a common mode current of 0.2 IxA amplitude per input. The measured interference, RTI, is about 30 IxV peal; and corresponds to an equivalent CMRR of 78.7 dB (16.3 dB decreased in view of Rp,~, RFB and CFB mismatch) CMRReqv = 20 log Fig. 6 Electrocardiogram (trace II) and intelference (trace I) acquired with proposed non-differential circuit trace shown in Fig. 6 is the OA U1A output, which can be used for power-line interference current monitoring. From its value (approximately 0.25 V pea];), the power-line interference current flowing through the electrodes, for this example, is 2 x 0.25 V /g = / . +/b - 3.3M~ / x ~/(2 x 3.14 x 5 0 H z x 2.2nF x 3.3 M ~ ) 2 + 1 ,-~ 0.4 btA peak The amplifier was tested in a laboratory environment with an oscilloscope supplied by the power-line through an uninterruptible power supply unit (UPS). The UPS power-line connecting contact plug was switched off during the measuring time. At the same time, the oscilloscope was supplied by the UPS battery and was isolated. Further, the circuit was tested without UPS isolation, and, in that case, the measured interference level was about twice as high. It should be noted that the limited value of the amplifier CMRR could be a reason for interference, but not the only one (WINTER and WEBSTER, 1983). In a two-electrode differential amplifier, the main components of the interference axe V~n ~ IcmAZ~ + IcmZcm/CMRR The CMRR and Zc,, axe responsible only for the second term of the above equation. Even with an infinite CMRR differential amplifier, the second term becomes zero, but the first term is able to produce significant noise proportional to Iv,, and AZe. The interference voltage shown in Figs 5 and 6 is about 100 mV peak. Referred to the amplifier input (RTI), it represents V#, = 0.2 mV peaR. The instrumentation amplifier AD620 has a specified minimum CMRR of 100 dB at a gain of 10 (AD620, Data Sheet). Using these values and ignoring the complex character of all the components, the approximate electrode imbalance AZe can be found, taking into consideration that Ic,, = Ig/2 = 0.2 IxA, Zc,,(50 Hz) = ZFB(50Hz) = 1.3 M f l (2.2 nFll3.3 Mfl) and CMRR = 105 (100 dB) AZ~ ~ Vin/Icm-Zcm/CMRR ,,~ l k - 1 3 ~ ,-~ lk. Additional measurements and experiments were performed. First, the input network capacitors CFB were decreased to 2 2 p F (this value ensures the interface stage stability and does not introduce a power-line frequency impedance Medical & Biological Engineering & Computing 2005, Vol. 43 IcmZFB V1N 0.21xA x 1 . 3 M ~ = 20 log = 78.7 dB 30pN In addition, an electrode imbalance of 1 k ~ was simulated by two 1 0 k ~ and 11 k ~ resistors connected in series to each input. The generator amplitude was increased to 4 mV to keep the same level of input current. The measured interference was approximately 0.25 mV peak RTI, and its value proves the above stated equation for Vin. It should also be noted that very precise matching of the input RC networks (ZFBa, ZFBb) is not needed, as the electrode impedances can differ dramatically by up to 100%. Therefore the off-the shelf component tolerances, e.g. 1% for resistors and 5% for capacitors, are quite acceptable. Nowadays, in modern biosignal instrumentation, the problem of reducing powerline noise becomes less critical owing to the presence of very efficient and powerful algorithms for its suppression and elimination (CHRISTOV and DOTSINSKY, 1988; DASKALOV et al., 1998; LEVKOV et al., 2005). Furthermore, removal of interference by software has no production cost. The amplifier described can be used for practically all kinds of two-electrode application. Other biosignal applications can be considered, depending on the types of electrode used. The moderately high differential input impedance of this amplifier would limit its application in cases where special, very high electrode-skin impedance electrodes are necessary and/or a high level of common mode interference has to be tolerated. In these cases, previously published designs are more convenient (DOBREV, 2002; 2004; DOBREV and DASKALOV, 2002). The following advantages of the proposed innovative amplifier circuit should be pointed out: 9 it is a very simple and cost effective solution, the component cost is less than $0.50 9 the simple, non-differential amplifier fully emulates a high CMRR differential by a tricky interface circuit 9 the amplifier is made from a small number of popular components; precise elements or matched resistors are not needed 9 the suggested amplifier principle eliminates the necessity of an instrumentation amplifier input stage in almost all kinds of two-electrode biosignal application. References CHRISTOV,I. I., and DOTSINSKY,I. A. (1988): 'New approach to the digital elimination of 50 Hz interference from the electrocardiogram', Med. Biol. Eng. Comput, 26, pp. 431 434 DASKALOV, I. K., DOTSINSKY, I. A., and CHRISTOV, I. I. (1998): 'Developments in ECG acquisition, preprocessing, parameter measurement and recording', IEEE Eng. ivied. Biol., 17, pp. 50 58 729 DOBREV, D. P. (2002): 'Two-electrode biopotential amplifier', Med. Biol. Eng. Comput., 40, pp. 546 549 DOBREV, D. P. (2004): 'Two-electrode low supply voltage electrocardiogram signal amplifier', Med. Biol. Eng. Comput., 42, pp. 272 276 DOBREV, D. P., and DASKALOV,I. K. (2002): 'Two-electrode biopotential amplifier with current-driven inputs', Med. Biol. Eng. Comput., 40, pp. 12 127 LEVKOV, CH., MIHOV, G., IVANOV, R., DASKALOV,I., CHRISTOV, I., and DOTSINSKY, I. (2005): 'Removal of power-line interference from the ECG: a review of the subtraction procedure', Biomed. Eng. Online, 4:50, http://www.biomedical-engineering-online.com/content/pdf/1475-925x-4-50.pdf NEUMAN, M. R. (1998): 'Biopotential amplifiers' in Webster, J. G. (Ed.): 'Medical instrumentation: applications and design, 3rd edn' (John Wiley & Sons, New York, 1998), pp. 262 264 THAKOR, N., and WEBSTER, J. (1980): 'Ground-free ECG recording with two electrodes', IEEE Trans. Biomed. Eng., 27, pp. 699 704 WINTER, B. B., and WEBSTER,J. G. (1983): 'Reduction of interference due to common mode voltage in biopotential amplifiers', IEEE Trans. Biomed. Eng., 30, pp. 58 62 AD620 Data Sheet, 9 Devices Inc., 2005, http:// www.analog.com HVC series Data Sheet, 9 OhmCraft Inc., 2005, http://www. ohmcraft.com Avnet price list, 2005, http://www.em.avnet.com 730 Authors" biographies DOBROMIRDOBREVobtained the MSc degree in Electronic Engineering from the Technical University of Sofia, in 1994. He received the PhD degree in 2000, in the field of Neonatal Monitoring. He has worked in the Institute of Medical Engineering of the Medical Academy and, since 1997, has been with the Centre of Biomedical Engineering, Bulgarian Academy of Sciences. He is currently an Analogue IC Designer at Melexis, Bulgaria, and his technical interests include analogue IC and systems design. TATYANANEYCHEVA obtained the MSc degree in Telecommunications from the Technical University of Sofia, in 1995. Her PhD, received in 2003, is on fast recovery defibrillator amplifiers. She is a research fellow at the Centre of Biomedical Engineering, Bulgarian Academy of Sciences, and her technical interests include digital signal processing and firmware development for embedded microcontroller systems. NIKOLAYMUDROVobtained the MSc degree in Electronic Engineering from the Technical University of Sofia, in 1973. His PhD, received in 1983, is on Pacemaker Signal Processing. He has worked at the Institute of Medical Engineering of the Medical Academy as a Senior Research Fellow, and since 1997, has been with the Centre of Biomedical Engineering, Bulgarian Academy of Sciences. His technical interests include electrical defibrillation, stimulation and biomedical signal processing. Medical & Biological Engineering & Computing 2005, Vol. 43